Method for processing an analog signal coming from a transmission channel, in particular a signal carried by power line communications

ABSTRACT

A method is for processing an analog signal coming from a transmission channel. The analog signal may include a useful signal modulated on a sub-set of carriers. The method may include analog-to-digital converting of the analog signal into a digital signal, and synchronization processing the digital signal. The synchronizing may include determining, in a time domain, a limited number of coefficients of a predictive filter from an autoregressive model of the digital signal, and filtering the digital signal in the time domain by a digital finite impulse response filter with coefficients based upon the limited number of coefficients to provide a filtered digital signal. The method may include detecting of an indication allowing a location in the frame structure to be identified, using the filtered digital signal and a reference signal.

RELATED APPLICATION

This application is based upon prior filed copending French ApplicationNo. 1552588 filed Mar. 27, 2015, the entire subject matter of which isincorporated herein by reference in its entirety.

TECHNICAL FIELD

The present disclosure is directed to transmission of information over acommunications channel and, in particular, to a transmission over anelectrical power line.

BACKGROUND

Power line communications technology aims to transmit digital data byexploiting the existing infrastructure of the electrical power grid. Itnotably allows the remote reading of electrical meters, the exchangesbetween electric vehicles, and the recharging terminals or else themanagement and the control power grids (smart grid). Power linecommunications (PLC) technology notably incorporates communication bynarrow-band power line communications (or N-PLC) which is generallydefined as a communication over an electrical power line operating attransmission frequencies up to 500 KHz.

N-PLC communication thus generally uses the bands of frequencies notablydefined by the European electrotechnical standards committee (ComitéEuropéen de Normalisation ELECtrotechnique-CENELEC) or by the FederalCommunications Commission (FCC). Thus, if the CENELEC A band offrequencies (3-95 kHz) is considered, the transmission frequencies aresituated between 42 and 89 KHz in the PRIME standard whereas they aresituated between 35 and 91 KHz for the PLC-G3 standard.

In these frequency bands, the electrical cables carrying the signals bypower line communications are in a very difficult environment. They arenotably subjected to interference of the white noise, colored noise, orpulse noise type. Furthermore, they are not protected against anyinterference. For this reason, any FM/AM radio signal or any wirelesscommunication can lead to the presence of harmonics of these signalswithin the useful frequency band used by narrow-band PLC communications.

Furthermore, the properties and characteristics of electrical powergrids are not known a priori and are variable over time. Thus,interference may be created on an electrical power line when a userconnects any given device such as for example a hair dryer or a washingmachine. This then results in a propagation of intense frequencyharmonics which may also be situated within the useful band of the PLCcommunications.

Accordingly, such noise signals, which are generally narrow-band noisesignals (i.e. Narrow Band Interferer), in other words having a smallerfrequency band than the frequency band of the useful signal, interferewith the synchronization phase of the receiver connected to theelectrical power line, during which the receiver must be able to besynchronized in order to notably locate the start of the useful data ofthe frame of symbols carried by the useful signal. The thesis by BrianMichael Donlan titled “Ultra-wideband Narrowband InterferenceCancellation and Channel Modeling for Communications”, 31 Jan. 2005,Blacksburg, Virginia, discloses various techniques for eliminatingnarrow-band noise signals from an ultra-wide-band (UWB) signal, inparticular, in the context of spectrum spreading (Spread Spectrum). Someof the approaches disclosed in this document use predictive filters soas to estimate the noise signal before subtracting it from the receivedsignal.

The signals mentioned in this document exhibit characteristics that arevery different from the signals used in the communications using powerline communications. Indeed, UWB signals (and, in particular, thoseusing direct sequence spread spectrum) exhibit a spreading of the powerof the transmitted signal over a wide band of frequencies in order tobury this power in the ambient noise or within the other communications.Thus, the power spectral density (or PSD) of a UWB signal is generallydefined as being less than −41 dBm/MHz.

The signals used in PLC communications are signals modulated accordingto a multi-carrier modulation, for example, a modulation in quadratureon orthogonal carriers (i.e. an Orthogonal Frequency DivisionMultiplexing (OFDM) modulation), but using only a sub-set of carriersfrom amongst a larger set of available carriers. Thus, for example, ifthe CENELEC A band of frequencies is considered, the size of the inverseFourier transform and of the direct Fourier transform is equal to 512,whereas only 97 sub-carriers (the sub-carriers 86 to 182) are used forthe transmission in the PRIME standard. If the CENELEC A band offrequencies is considered, the size of the inverse Fourier transform andof the direct Fourier transform is equal to 256 whereas only 36sub-carriers (the sub-carriers 23 to 58) are used in the PLC-G3standard. Furthermore, it may be useful, during the synchronizationphase, not to miss any symbol coming from the channel even when thelatter is affected by noise.

SUMMARY

Generally speaking, a method is for processing an analog signal comingfrom a transmission channel. The analog signal may include a usefulsignal modulated on a sub-set of carriers from a plurality of availablecarriers. The useful signal carries at least one frame of symbolsaccording to a frame structure, the useful signal to be affected bynoise from at least one noise signal in a narrow band. The method mayinclude analog-to-digital converting of the analog signal into a digitalsignal, and synchronization processing the digital signal. Thesynchronizing may include determining, in a time domain, a limitednumber of coefficients of a predictive filter from an autoregressivemodel of the digital signal, and filtering the digital signal in thetime domain by a digital finite impulse response filter withcoefficients based upon the limited number of coefficients to provide afiltered digital signal. The method may include detecting of at leastone indication allowing at least one location in the frame structure tobe identified, using the filtered digital signal and a reference signal.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic diagram of a transmitter, according to the presentdisclosure.

FIG. 2 is a schematic diagram of modulated carriers form a sub-set SNSof carriers, according to the present disclosure.

FIG. 3 is a schematic diagram of a receiver, according to the presentdisclosure.

FIG. 4 is a diagram of a frequency spectrum, according to the presentdisclosure.

FIG. 5 is a schematic diagram of a frame of data, according to thepresent disclosure.

FIG. 6 is a diagram of a frequency response of a filter, according tothe present disclosure.

FIG. 7 is a flowchart of a filtering process, according to the presentdisclosure.

FIG. 8 is a schematic diagram of a filter, according to the presentdisclosure.

FIG. 9 is a flowchart of a filtering process, according to the presentdisclosure.

FIG. 10 is a schematic diagram of a processor, according to the presentdisclosure.

DETAILED DESCRIPTION

Various embodiments of the present disclosure and their implementationare compatible with the various standards governing communication bypower line communications, notably but not exclusively the standardsPLC-G3, PoweRline Intelligent Metering Evolution (PRIME) or else thestandard IEEE 1901-2. According to one embodiment and itsimplementation, a processing of an analog signal coming from thetransmission channel is provided which allows the performancecharacteristics of the synchronization phase to be improved whether theanalog useful signal is affected by noise or not from at least onenarrow-band noise signal. According to another embodiment and itsimplementation, the aim is also to improve the performancecharacteristics of the decoding of the remaining part of the frame aftersynchronization.

According to one embodiment and its implementation, in order tofacilitate the synchronization of the receiver, notably when the usefulsignal is affected by noise from at least one narrow-band noise signal,coefficients of a predictive filter are determined on the fly in thetime domain from an autoregressive model of the signal and the signal inthe time domain is filtered on the fly by a digital finite impulseresponse (or FIR) filter whose coefficients are those of the predictivefilter.

This therefore differs from the filtering methods of the prior art inwhich the noise signal is estimated prior to subtracting it from thesignal coming from the channel, which requires a perfectsynchronization, both in time and in phase, and which is difficult toachieve in practice, notably in the case of a phase synchronizationwhere it is important, on the one hand, not to lose any symbol and, onthe other hand, not to introduce phase errors.

According to this embodiment and its implementation, instead ofestimating then of subtracting the estimated noise signal, the overallsignal coming from the channel is filtered, whether this signal containsor does not contain the noise signal, which allows a non-coherentprocessing to be carried out, in other words not requiring a perfectsynchronization in time and in phase between the moment when the noisesignal is estimated and the moment when this noise signal is subtractedfrom the global signal. The receiver will thus be able to besynchronized using the filtered signal and a reference signal, forexample a known symbol.

According to one aspect, a method is therefore provided for processing achannel analog signal coming from a transmission channel, for example anelectrical power line. The channel analog signal is able to comprise auseful signal modulated on a sub-set from a set of available carriers,such as for example the useful signals conforming to the PRIME or G3-PLCstandards. This useful signal carries at least one frame of symbolsaccording to a frame structure and this useful signal could be affectedby noise from at least one narrow-band noise signal.

In general, the noise signal is a noise peak at a single frequencycontained within the band of frequencies of the useful signal but, moregenerally, a narrow-band noise signal is a noise signal whose band offrequencies is smaller than the band of frequencies of the usefulsignal. Thus, the channel analog signal may perfectly well, at a givenmoment in time, not comprise any useful signal or else only comprise atleast one noise signal, or else a noise-free useful signal or a usefulsignal affected by noise.

The method according to this aspect then comprises an analog-digitalconversion of the channel analog signal and a synchronization processingcomprising a filtering processing. The channel analog signal that willundergo the analog/digital conversion may for example be the analogsignal coming directly from the channel or else, as is generally thecase, the analog signal delivered by an analog input stage (notablycomprising band pass filters, low-pass filters and an amplifier)connected to the transmission channel.

The filtering processing includes a determination on the fly in the timedomain of a limited number of coefficients of a predictive filter froman autoregressive model of a channel digital signal coming from theanalog/digital conversion and a filtering on the fly of the channeldigital signal in the time domain by a digital finite impulse responsefilter whose coefficients are those of the predictive filter. Thechannel digital signal on which the filtering processing is carried outis not necessarily the digital signal coming directly from theanalog/digital conversion but may for example be the digital signalcoming from the analog/digital conversion and which could have beensubjected to an under-sampling.

The signals modulated on a sub-set of carriers from amongst a set ofavailable carriers have totally different characteristics from the UWBsignals or with spread frequency spectrum. They do indeed notablyexhibit a level of power much higher than that of a UWB signal or signalwith spread frequency spectrum and it is then preferable to takeprecautions in the filtering so as to avoid completely filtering theuseful signal in the absence of narrow-band noise.

Indeed, in view notably of the fact that only a few carriers are usedfor the modulation from amongst the available set of carriers (size ofthe inverse or direct Fourier transform), the number of coefficients ofthe filter should preferably be limited so as to avoid having too highof an attenuation of the useful signal in the absence of a noise signal.

In other words, the number of coefficients of the filter isadvantageously less than or equal to a limiting number which is chosenso as to form a finite impulse response filter whose frequency responsecomprises, in the presence of the noise signal, a notch at the frequencyband of the noise signal, and whose frequency response, in the absenceof a noise signal, has a relatively flat profile in the band offrequencies of the useful signal in such a manner as to allow anattenuation of the useful signal less than a chosen value, for example 6dB, which does of course depend on the application envisioned.

The number of coefficients will furthermore preferably be equal to thislimiting number so as to more easily take into account potentiallyseveral narrow-band noise signals with different tones. Those skilled inthe art will be able to adapt the limiting number of coefficientsaccording to the application envisioned and the desired characteristicsfor the maximum acceptable attenuation of the useful signal in theabsence of a noise signal. However, the inventors have observed that,when each symbol comprises a cyclical prefix, the acceptable limitingnumber for coefficients of the filter is around three quarters, andpreferably around half of the length of the cyclical prefix expressed innumber of samples.

It is recalled here that the cyclical prefix mainly allows theinter-symbol interferences to be eliminated and is a technique whichconsists in copying one part of a symbol to place it upstream of thissymbol. In practice, the poles of the predictive filter coming from theautoregressive model of the signal become the zeros of the finiteimpulse response filter. Accordingly, the FIR filter can only attenuatethe frequencies corresponding to these zeros.

So, in such a manner as to further limit the attenuation of the usefulsignal during the filtering, it is advantageous for the coefficients ofthe predictive filter to be calculated at a processing frequency in therange between 2 and 5 times, and preferably between 2 and 3 times, themaximum frequency of the channel digital signal. This allows the usefulsignal to then be considered for the filter as a kind of white noisewith respect to the noise signal, which would not be the case if theprocessing frequency were much higher than the maximum frequency of thechannel digital signal.

The determination of the coefficients of the filter and the filteringitself are carried out in the time domain and on the fly, in other wordsprogressively with the arrival of the channel analog signal. This avoidsany symbol being missed whether the signal is affected by noise or not.The method according to this aspect also comprises a detection of atleast one indication allowing at least one location in the framestructure to be identified, using the filtered channel digital signaland a reference signal. The indication may for example be therecognition of a known symbol from the preamble sequence of a frame andthe reference signal may be this known symbol, the detection then beingapplied for example by moving correlation operations.

According to one embodiment, the determination on the fly of thecoefficients and the filtering on the fly comprise an organization ofthe samples of the digital signal into successive groups of samples, adetermination of a current block of coefficients using the current groupof samples, and an application to the current group of the finiteimpulse response filter having the current block of coefficients in sucha manner as to obtain a filtered current group of samples.

According to one embodiment, each frame comprises a preamble sequencecomprising known symbols and preceding the remaining part of the frame,and the filtering processing is applied at least for detecting the atleast one indication within the preamble sequence of at least one frame.When the transmission is totally asynchronous, in other words thereceiver does not know the time period between successive frames, it isadvantageous for the filtering processing operations to be applied atleast for detecting the indication on the preamble sequence of eachframe.

For the filtered digital signal, it cannot be known whether thisfiltered signal results from a useful signal affected by noise or from anoise-free useful signal. It is accordingly advantageous, after thedetection of the indication, to carry out a verification of the presenceor of the absence of the noise signal, for example based on at least oneknown symbol of the unfiltered useful signal. Indeed, this will allowthe processing of the further symbols of the frame to be improved.

This verification may comprise a direct Fourier transform processing onthe unfiltered known symbol and an analysis of the power of eachcarrier. This verification is for example carried out on a symbol of thepreamble sequence. Then, in the case of absence of the noise signal, theprocessing of the remaining part of the frame is advantageously carriedout on the unfiltered channel digital signal, which enables the decodingof the symbols of the remaining part of the frame to be carried out onan unfiltered, in other words unattenuated, signal. On the other hand,in the presence of the noise signal, the processing of the remainingpart of the frame will be carried out on the filtered channel digitalsignal.

This processing of the remaining part comprises a direct Fouriertransform processing, a de-mapping processing supplying for each carriera value of the modulation coefficient (each symbol comprises modulationcoefficients, or “bins”, respectively associated with the carriers) anda determination, for each modulation coefficient, of a confidenceindicator (soft decision) for the value. It is then advantageous toforce to zero the confidence indicators of the modulation coefficientsassociated with the carriers whose frequencies correspond to those ofthe noise signal. This allows the performance characteristics of thedecoding to be improved notably when a de-interlacer-decoder pair of theViterbi type (or any other decoder based on soft decisions) is used. Itis also preferable, in order not to interfere too much with thedecoding, to freeze the values of the coefficients of the filter duringthe processing of the remaining part of the frame.

According to another aspect, a receiver is provided, comprising an inputstage designed to be connected to a transmission channel and configuredfor delivering a channel analog signal coming from the transmissionchannel, the channel analog signal being able to comprise a usefulsignal modulated on a sub-set from a set of available carriers, carryingat least one frame of symbols according to a frame structure and whichcould be affected by noise from at least one narrow-band noise signal,an analog/digital conversion stage for performing an analog/digitalconversion of the channel analog signal, and a processing stagecomprising filtering means or a filter including a calculation moduleconfigured for determining, on the fly, a limited number of coefficientsfor a predictive filter from an autoregressive model of a channeldigital signal coming from the analog/digital conversion stage, adigital finite impulse response filter, whose coefficients are those ofthe predictive filter, for performing a filtering on the fly of thechannel digital signal in the time domain and detection means or adetector configured for detecting at least one indication allowing atleast one location in the frame structure to be identified, using thefiltered digital signal and a reference signal.

According to one embodiment, each symbol comprises a cyclical prefix andthe number of coefficients of the filter is less than or equal to alimited number which is of the order of three quarters, preferably ofthe order of a half, of the length of the cyclical prefix expressed innumber of samples. According to one embodiment, the calculation moduleis configured for calculating the coefficients of the predictive filterat a processing frequency in the range between 2 and 5 times, preferablybetween 2 and 3 times, the maximum frequency of the channel digitalsignal.

According to one embodiment, the processing stage comprises organizingmeans or an organizer circuit configured for regrouping the samples ofthe channel digital signal into successive groups of samples, thecalculation module is configured for determining a current block ofcoefficients using the current group of samples, and the digital filteris configured for receiving the current group of samples at the input soas to deliver a filtered current group of samples.

According to one embodiment, each frame comprises a preamble sequencecomprising known symbols and preceding the remaining part of the frame,and the processing stage comprises control means or a controllerconfigured for delivering the channel digital signal to the filteringmeans at least so that the detection means detect the at least oneindication within the preamble sequence of at least one frame. Accordingto one embodiment, the control means are configured for delivering thechannel digital signal to the filtering means at least so that thedetection means detect the at least one indication within the preamblesequence of each frame.

According to one embodiment, the receiver furthermore comprisesverification means or a verification circuit configured for carryingout, after the detection of the at least one indication, a verificationof the presence or of the absence of the noise signal based on at leastone symbol of the unfiltered useful signal. According to one embodiment,the verification means comprise a direct Fourier transform stageconfigured for performing a direct Fourier transform processing on theat least one symbol and means of analysis configured for carrying out ananalysis of the power of each carrier.

According to one embodiment, the verification means are configured forcarrying out the verification on at least one symbol from the preamblesequence. According to one embodiment, the processing stage furthermorecomprises additional processing means or an additional processorconfigured for carrying out a processing of the remaining part of eachframe and in the case of the absence of the noise signal, the controlmeans are configured for delivering the remaining part of the framedirectly to the additional processing means without going through thefiltering means.

According to one embodiment, each symbol comprising modulationcoefficients respectively associated with the carriers, the additionalprocessing means comprise a direct Fourier transform stage, a de-mappingmeans or a de-mapper supplying for each carrier a value of themodulation coefficient and a module able to determine a confidenceindicator of the value for each modulation coefficient, and forcingmeans or zeroing circuit configured for, in the case of presence of thenoise signal, forcing to zero the confidence indicators for themodulation coefficients associated with the carriers whose frequenciescorrespond to those of the noise signal.

According to one embodiment, the control means are configured fordisabling the module for calculating the coefficients of the filterduring the processing of the remaining part of the frame (freezing ofthe coefficients of the filter). According to one embodiment, the usefulsignal is a signal modulated according to an OFDM modulation.Furthermore, the transmission channel can be an electrical power line,the useful signal then being a signal carried by power linecommunications.

According to yet another aspect, independently of an application to asynchronization, a method is provided for filtering a channel analogsignal coming from a transmission channel, the channel analog signalbeing able to comprise a useful signal, for example modulated on asub-set from a set of available carriers, and which could be affected bynoise from at least one narrow-band noise signal, the method comprisingan analog/digital conversion of the channel analog signal and afiltering processing including a determination on the fly in the timedomain of coefficients of a predictive filter from an autoregressivemodel of a channel digital signal coming from the analog/digitalconversion and a filtering on the fly of the channel digital signal inthe time domain by a digital finite impulse response filter whosecoefficients are those of the predictive filter. The number ofcoefficients is advantageously limited as indicated hereinbefore and/orthe coefficients of the predictive filter are calculated at a processingfrequency in the range between 2 and 5 times, preferably between 2 and 3times, the maximum frequency of the channel digital signal. According toyet another aspect, a receiver is provided designed to be connected tothe transmission channel and comprising means or circuitry, such asthose defined hereinbefore, configured for implementing such a filteringmethod.

The embodiments and their implementation will now be described withinthe framework of a transmission of information via power linecommunications (PLC), although the present disclosure is not limited tothis type of application. In the following part of the description, eachtime that the standards PLC-G3 or PRIME are mentioned by way ofnon-limiting examples, it will be assumed that the CENELEC A band offrequencies (3-95 kHz) are being considered.

Reference is now made to FIG. 1, a transmitter is capable oftransmitting a useful signal SU over an electrical power line LE bypower line communications. The transmission chain comprises, forexample, an encoder ENC (e.g. a convolution-type encoder) receiving thedata to be transmitted from source coding means or a source code store.Interlacing means or a interlacing circuit INTL are connected to theoutput of the encoder and are followed by “mapping” means or a mapperwhich transform the bits into symbols according to a transformationscheme depending on the type of modulation used, for example amodulation of the BPSK type or, more generally, a QAM modulation. Eachsymbol contains modulation coefficients associated with carriers whichwill be modulated accordingly. The symbols are delivered at the input ofprocessing means or a processor MTFI designed to perform an inverse fastFourier transform (IFFT) operation.

Referring now to FIG. 2, it will be noted that the modulated carriersform a sub-set SNS of carriers from amongst a set ENS of availablecarriers (a set which corresponds to the size of the inverse Fouriertransform). Thus, in the PLC-G3 standard, the size of the inverseFourier transform is equal to 256, whereas the modulated carriers of thesub-set SNS are included between the ranks 23 and 58, which correspondsto a frequency band F1-F2 in the range between 35 and 91 KHz. Thesampling frequency here is equal to 400 KHz leading to a separationbetween the carriers equal to 1.5625 KHz, which thus renders thefrequencies orthogonal (i.e. the OFDM modulation).

In the PRIME standard, the size of the inverse Fourier transform isequal to 512, whereas the number of carriers of the sub-set SNS is equalto 97, which provides a frequency band extending between 42 and 89 KHzfor the useful signal. The modulation coefficients associated with theunused carriers are equal to 0.

The OFDM signal in the time domain is generated at the output of theprocessing means MTFI, and means or an adder circuit MCP add to eachOFDM symbol in the time domain, a cyclical prefix which is a copy, inthe header of the OFDM symbol, of a certain number of samples situatedat the end of this symbol. By way of example, in the PLC-G3 standard,the length of the cyclical prefix is 30 samples for a sampling frequencyof 400 KHz, whereas it is 48 samples for a sampling frequency of 250 KHzin the PRIME standard. The signal is subsequently converted in adigital/analog converter CNA then processed in a stage ETA, commonlydenoted by those skilled in the art using the term “Analog Front End”,where it notably undergoes a power amplification, prior to beingtransmitted over the electrical power line LE.

In reception, referring now to FIG. 3, it can be seen that the receiverRCP here comprises an analog input stage ET1 whose input terminal BE isconnected to the electrical power line LE. The analog input stage ET1typically comprises a band pass filter BPF, a low-pass filter LPF, andmeans of amplification or an amplifier AMP. The output of the stage ET1is connected to an analog/digital conversion stage CAN whose output isconnected to the input of a processing stage ET2. The processing stageET2 here comprises automatic gain control means or an automatic gaincontrol circuit AGC allowing the value of the gain of the amplificationmeans AMP of the stage ET1 to be controlled.

The signal SAC delivered at the output of the analog stage ET1 and atthe input of the analog/digital conversion stage CAN denotes a channelanalog signal (i.e. an analog signal) coming from the transmissionchannel (electrical power line) LE. By way of non-limiting example, thefrequency spectrum of such a channel analog signal SAC is illustratedschematically in FIG. 4.

It can be seen that this signal SAC comprises the useful signal SUcarrying the data transmitted from the transmitter and whose band offrequencies is situated between the frequencies F1 and F2 correspondingto the numbers of the modulated carriers. The signal SAC alsopotentially comprises a narrow-band noise signal SB, which willpotentially interfere with the useful signal SU.

Generally, the noise signal SB comprises a single tone situated at thefrequency F3. It may however in practice be distributed over the carrierwith a frequency F3 and also over a few adjacent carriers. It can beseen that the signal SU is a signal in the shape of a dome whose levelis much higher than the level of AWGN noise of the channel in theabsence of a signal. The level of the noise signal SB is on the otherhand higher than the level of the useful signal SU.

Referring now back to FIG. 3, it can be seen that the processing stageET2 also comprises a low-pass filter LPF2 followed, although this is notindispensible, by under-sampling means or an under-sampler MSCH. Thesampling frequency of the signal upstream of the means MSCH is denotedFs, whereas the sampling frequency of the signal at the output of themeans MSCH is denoted Fss.

The signal SNC at the output of the means MSCH then denotes here achannel digital signal which is coming from the analog/digitalconversion of the channel analog signal SAC and to which a filteringprocessing in filtering means or a filter MSL will notably be applied aswill be seen in more detail hereinafter.

In the following part, the frequency Fc denotes the processing frequencyat which will notably be calculated the coefficient of the filter of thefiltering means MFL. In the G3-PLC standard, for example, the samplingfrequency Fs specified is 400 KHz for a size of FFT of 256.

Although it would have been possible to carry out all the filteringoperations at a processing frequency Fc equal to the sampling frequencyFs of 400 KHz, under-sampling the signal at a frequency Fss less thanFs, and carrying out all the filtering operations at the processingfrequency Fc equal to Fss allows the complexity of implementation of theprocessing stage, and notably of the filtering means, to be reduced andalso allows a direct fast Fourier transform (FFT) processing to becarried out with a reduced size with respect to the specified size of256.

Before returning in more detail to the structure of the filtering meansMFL and to the other means that are incorporated into the processingstage ET2, reference is now made to FIG. 5 in order to illustrate thestructure of a frame carrying symbols, for example in the framework ofthe PLC-G3 standard. The frame TRM comprises a preamble sequence PRMhere comprising eight known symbols SYNCP followed by a symbol ofopposing phase SYNCM itself followed by a semi-symbol SYNCM.

The frame TRM subsequently comprises a header (preamble sequence) HDfollowed by a field PLD containing useful data symbols to be decoded andbetter known by those skilled in the art using the term “payload”. Thesymbols of the header HD notably contain control information for thedecoding of the data in the field PLD together with the number of bytesto be decoder in the field PLD.

The preamble sequence PRM of the frame TRM allows the receiver to besynchronized, in other words an indication to be obtained allowing thestructure of the frame to be recovered in order to be able to identifythe start of the header HD. The filtering means MFL will be applied, atleast during the synchronization phase of the receiver and potentially,as will be seen in more detail hereinafter, during the phase fordecoding the remaining part of the frame TRM (header and field PLD) inthe case where a noise signal proves to be present.

The filtering means MFL will determine, on the fly, coefficients of apredictive filter from an autoregressive model of the channel digitalsignal SNC and will then filter, on the fly, the channel digital signalin the time domain by a digital finite impulse response filter whosecoefficients are those of the predictive filter.

As is well known by those skilled in the art, a signal may be modeled bymeans of a white noise convoluted with an autoregressive filter. Theparameters of the model (the coefficients of the predictive filter andthe variance of the prediction error) can be estimated based on theself-covariance of the signal by solving the Yule Walker equations:

${{R_{n + 1}A_{n}} = {{\begin{bmatrix}r_{0} & r_{1}^{*} & \ldots & r_{n}^{*} \\r_{1} & \ddots & \ddots & \vdots \\\vdots & \ddots & \ddots & r_{1}^{*} \\r_{n} & \ldots & r_{1} & r_{0}\end{bmatrix}\begin{bmatrix}1 \\A_{n,1} \\\vdots \\A_{n,n}\end{bmatrix}} = \begin{bmatrix}\sigma_{f,n}^{2} \\0 \\\vdots \\0\end{bmatrix}}};$

in which

$A_{n} = \begin{bmatrix}1 \\A_{n,1} \\\vdots \\A_{n,n}\end{bmatrix}$

are the n coefficients of the predictive filter of the autoregressivemodel of order n and σ_(f,n) ² is the variance of the prediction error.The sign * denotes the complex conjugate.

The self-covariance sequence R_(n+1)=[r₀ r₁ . . . r_(n)] may beestimated by the following formula:

$r_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1 - k}{y_{n + k}y_{n}}}}$k = 0, 1, …  , n;

in which y is a sequence of N samples of the input signal.

Generally speaking, N must be sufficiently big in order to include allthe periodic content of the signal and in order to render random anynon-periodic content. However, in practice, N can be equal to the sizeof the symbol, potentially under-sampled, which also corresponds to thesize of the Fourier transform. Several algorithms exist for solving theYule Walker equations. The Levinson algorithm may notably be mentioned,or the Durbin-Watson algorithm or else the Burg algorithm oralternatively an algorithm of the least squares type. Those skilled inthe art will notably be able to refer, for this purpose, to page 879 ofAppendix A of the book by John G. Proakis, 3^(rd) edition, entitledDigital Communications or else to Chapters 11-4-2 and 11-1-2 of thissame book.

When the Levinson algorithm is used, the latter is a recursive algorithmthat calculates the coefficients one by one for example according to thefollowing sequence:

A ₀=[1]; σ_(f,0) ² =r ₀

-   -   repeat for m=0 to n:

$\Delta_{m + 1} = {\begin{bmatrix}r_{m + 1} & \ldots & r_{1}\end{bmatrix}A_{m}}$$K_{m + 1} = \lbrack \frac{- \Delta_{m + 1}}{\sigma_{f,m}^{2}} \rbrack$$A_{m + 1} = {\begin{bmatrix}A_{m} \\0\end{bmatrix} + {K_{m + 1}\begin{bmatrix}0 \\{JA}_{m}^{*}\end{bmatrix}}}$ σ_(f, m + 1)² = σ_(f, m)²(1 − K_(m + 1)²).

Once the coefficients An of the predictive filter are determined, afinite impulse response filter (FIR filter) is then constructed whosetransfer function in z is defined by the formula hereinbelow:

1+A ₁ z ⁻¹ +A ₂ z ⁻² +A ₃ z ⁻³+ . . . .

In this formula, the coefficients An of the FIR filter are thecoefficients An of the predictive filter of the aforementionedautoregressive model. The set of coefficients of the filter isadvantageously limited, in other words less than or equal to a limitingnumber and preferably equal to this limiting number. Indeed, in viewnotably of the fact that the useful signal is modulated on only asub-set of carriers from a set of available carriers, too large a numberof coefficients of the filter would run the risk of causing too high anattenuation of the signal during the filtering, in particular when thereis no noise signal.

Generally speaking, the limited number of coefficients is chosen bythose skilled in the art, taking into account the application and thespecifications envisioned, so that, as illustrated schematically in FIG.6, the frequency response Hl of the filter in the presence of anarrow-band noise signal has a notch in the neighborhood of thefrequency F3 of the noise signal, and so that the frequency response H2of this filter exhibits, in the absence of the noise signal, arelatively flat profile in the frequency band F1, F2 of the usefulsignal in such a manner as to obtain an attenuation of the signal lowerthan an acceptable limiting attenuation.

This acceptable limiting attenuation depends on the implementation andon the dynamic behavior supported by the various processing means orprocessors. Those skilled in the art will be able to choose thisacceptable limiting attenuation as a function of these conditions.However, by way of non-limiting example, the acceptable limitingattenuation can be of the order of 6 dB.

In practice, in order to satisfy this condition, a number ofcoefficients for the filter could, for example, be chosen that is lessthan or equal to three quarters, and preferably to a half, of the lengthof the cyclical prefix expressed in number of samples taking intoaccount the processing frequency Fc used. In the PLC-G3 standard, thelength of the cyclical prefix for a sampling frequency Fs of 400 KHz is30 samples. Thus, if the PLC-G3 standard is considered, a number ofcoefficients equal to 15 Fc/Fs could for example be chosen.

In practice, as illustrated in FIG. 7, the filtering processing carriedout on the fly in the time domain includes a grouping of samples (step70) so as to form a current group GR of N samples. Then, in the step 71,the coefficients of the predictive filter are calculated by executingthe Levinson algorithm in accordance with the aforementioned sequencefor m varying from 0 to the limiting value for the number ofcoefficients. Then, in the step 72, the current group GR of N samples inthe time domain is filtered with the finite impulse response filterwhose coefficients are those that have just been calculated for thepredictive filter. A group GRF of N filtered samples is then obtained.

Furthermore, it is advantageous for the processing frequency Fc of thecoefficients of the filter (equal to the frequency Fs or potentially tothe frequency Fss in the case of under-sampling) not to be too high withrespect to the maximum frequency of the channel digital signal(potentially under-sampled). Indeed, if the processing frequency Fc istoo high with respect to this maximum frequency, the channel digitalsignal will not be seen as a “white” noise with respect to the noisesignal and there is a risk of having too high an attenuation of theuseful signal. Thus, a processing frequency Fc may be chosen in therange between 2 and 5 times, and preferably between 2 and 3 times, themaximum frequency of the channel digital signal on which the coefficientof the filter is calculated.

In practice, as illustrated in FIG. 8, the filtering means MFLfunctionally comprise means or circuitry MGR configured for organizingthe samples into groups of samples, a module MCL for calculatingcoefficients of the predictive filter and a module FIR implementing thefinite impulse response filter. In practice, these various circuits andmodules can, for example, be implemented by software within amicroprocessor.

If reference is again made to FIG. 3, it can be seen that the filtereddigital signal SNF delivered by the filtering means MFL is notably usedby synchronization means or a synchronizer MSYNC, with a conventionalstructure known per se, in order to allow the receiver RCP to besynchronized, in other words, for example, to identify the structure ofthe frame and its timing in such a manner as to be able to correctlydecode the header HD and the field PLD.

More precisely, the synchronization means MSYNC perform movingcorrelation processing operations between the filtered digital signalSNF and a reference signal SREF which is, in the present case, a knownsymbol of the frame, for example, a known symbol of the preamblesequence, such as the symbol SYNCP. In the example described here, theindication IND representative of the frame structure and of an appliedsynchronization will, for example, be the occurrence of the transitionbetween the last symbol SYNCP of the preamble sequence and the symbolSYNCM.

This indication IND will be transmitted to the additional processingmeans MTRS of the processing stage ET2 so as to allow the decoding ofthe symbols of the header HD and of the field PLD of the frame. However,by simply observing the filtered digital signal, it is very difficult,or even impossible, to know whether this filtered digital signal resultsfrom a useful signal affected by noise or from a noise-free usefulsignal.

Nevertheless, it is advantageous, as will be seen in more detailhereinafter, to know this information so as to further improve theperformance characteristics of the decoding of the remaining part of theframe. For this purpose, the processing stage ET2 comprises verificationmeans or a verification circuit MVRF configured for verifying thepresence or otherwise of the noise signal within the useful signal, oncethe synchronization has been carried out. More precisely, thisverification will be carried out on the preamble sequence of theunfiltered channel digital signal SNC, and more particularly, on one ofthe symbols of the preamble sequence, for example the unfiltered symbolSYNCP.

As illustrated in FIG. 9, a direct fast Fourier transform FFT is appliedin a step 86 in such a manner as to carry out a transformation from thetime domain to the frequency domain, then (in step 91) a power analysisis carried out on the lines of the frequency spectrum obtained at theoutput of the Fourier transform. For this purpose, in the step 92, it isexamined whether certain frequency lines have a power or a level greaterthan a fixed threshold TH.

If none of the frequency lines of the spectrum has a level higher thanthe threshold TH, then it is concluded that there is an absence of anoise signal SB within the channel digital signal SNC. In the oppositecase, if at least one line has a power higher than the threshold TH,then the presence of a narrow-band noise signal SB is concluded. Inaddition, this analysis also allows the position in the frequencyspectrum of the noise signal to be known, in other words which are thebins concerned.

Accordingly, as illustrated schematically in FIG. 3, the verificationmeans MVRF comprise means or a direct Fourier transform circuit MTFDconfigured for performing the direct Fourier transform processing,together with means of analysis or an analyzer circuit MAL. In practice,these means may here again be implemented for example by software withina microprocessor. Furthermore, as will be seen hereinafter, the meansMTFD are advantageously those which are already present within theadditional processing means MTRS.

For the following part of the processing, in other words the decoding ofthe symbols of the remaining part of the frame, the latter will becarried out on the unfiltered channel digital signal SNC if the resultof the aforementioned verification is that this signal SNC was not infact affected by noise from the narrow-band noise signal SB. On theother hand, if the verification shows that the noise signal was present,then the processing of the remaining part of the frame will continue tobe carried out on the filtered digital signal SNF delivered by thefiltering means MFL.

For this purpose, control means or a controller, here corresponding byway of illustration to a multiplexer MUX controlled by a signal SCdelivered by a control module MC connected to the output of theverification means MVRF and representative of the presence or of theabsence of the noise signal, will enable or disable the filtering meansMFL for the subsequent processing of the remaining part of the frame.The control module may for example be formed by a logic circuit or elsebe implemented by software. More precisely, as illustrated schematicallyin FIG. 3, in the case of absence of a noise signal, the channel digitalsignal SNC is delivered directly to the additional processing meansMTRS, whereas in the presence of the noise signal SB, it is the filtereddigital signal SNF that is delivered to the additional processing meansMTRS.

If reference is now made to FIG. 10, it can be seen that thesecomplementary processing means MTRS comprise means or an extractorcircuit MCPR configured for extracting the cyclical prefix from eachsymbol, followed by the means MTFD configured for carrying out thedirect fast Fourier transform FFT. The means MTFD are followed byde-mapping means or a de-mapper DMP supplying, for each carrier, acorresponding value of the modulation coefficient (bin). Thesede-mapping means DMP are followed by a module MCSM configured fordetermining, for each modulation coefficient, a confidence indicator(“soft decision”) for the value. This module is conventional and knownper se and uses for example an algorithm of the LogMAP type.

The additional processing means MTRS also comprise de-interlacing meansor a de-interlacing circuit DINTL followed by a decoder DCD, forexample, a decoder of the Viterbi type, followed by means or a cyclicredundancy check (CRC) circuit CRC able to carry out a parity check. Theoutput of the means CRC is connected to the output terminal BS of themeans MTRS which is connected to the means forming the MAC layer of thereceiver. When the additional processing means MTRS receive at the inputthe filtered digital signal, in other words in the presence of a noisesignal on the channel digital signal SNC, it is advantageous for theconfidence indicators (soft decision) associated with the bins on whichthe noise signal is present, together with those potentially associatedwith the neighboring bins, to be set to zero. Indeed, such zero softdecisions are seen as being neutral decisions for the error correctionalgorithm implemented in the Viterbi decoder.

This allows the performance of the decoding of the de-interlacer-decoderpair to be further improved. For this purpose, the processing means MTRFcomprise forcing means or a zeroing circuit MFC configured for applyingthis forcing to zero. Here again, in practice, all of these means andmodules of the additional processing means MTRS may be implemented bysoftware modules within a microprocessor. Furthermore, so as to notinterfere with the decoding of the remaining part of the frame when thefiltering means MFL are enabled, it is preferable, for the decoding ofthis remaining part of the frame, for the coefficients of the FIR filterto be fixed, in other words not to progressively update them along withthe decoding of the remaining part of the frame. For this purpose, thecontrol module MC can deliver a signal SC1 to the calculation module MCLfor freezing the coefficients of the filter.

According to one aspect of the present disclosure, it is thus possibleto obtain a notable improvement in performance in the synchronizationphases and in the decoding of the symbols of the frame notably in thepresence of narrow-band noise signals, which have levels that can go upto 50 to 60 dB above the level of the useful OFDM signal, whereas thecurrent standards only require a robust decoding in the presence of anoise signal whose level only exceeds that of the useful signal by 20dB.

1-28. (canceled)
 29. A method for processing an analog signal from atransmission channel, the analog signal comprising a useful signalmodulated on a sub-set of carriers from a plurality of availablecarriers, the useful signal carrying at least one frame of symbolsaccording to a frame structure, the useful signal susceptible to noise,the method comprising: analog-to-digital converting of the analog signalinto a digital signal; synchronizing the digital signal, thesynchronizing comprising determining, in a time domain, a number ofcoefficients of a predictive filter from an autoregressive model of thedigital signal, and filtering the digital signal in the time domain by adigital finite impulse response filter with coefficients based upon thenumber of coefficients to provide a filtered digital signal; anddetecting at least one indication for at least one location in the framestructure to be identified and based upon the filtered digital signaland a reference signal.
 30. The method according to claim 29 whereineach symbol comprises a cyclical prefix; and wherein a number of thecoefficients of the filtering of the digital signal is less than orequal to half of a sample length of the cyclical prefix.
 31. The methodaccording to claim 29 wherein the coefficients of the predictive filterare calculated at a processing frequency in a range between 2 and 5times of a threshold frequency of the digital signal.
 32. The methodaccording to claim 29 wherein the determining of the coefficients andthe filtering comprise: organizing of samples of the digital signal intosuccessive groups of samples; determining a current block ofcoefficients using a current group of samples; and applying to thecurrent group of the finite impulse response filter having the currentblock of coefficients so as to obtain a filtered current group ofsamples.
 33. The method according to claim 29 wherein each framecomprises a preamble sequence comprising known symbols, and a remainingpart subsequent to the preamble sequence; and wherein the filtering isapplied at least for detecting the at least one indication within thepreamble sequence of at least one frame.
 34. The method according toclaim 33 wherein the filtering is applied at least for detecting the atleast one indication within the preamble sequence of each and everyframe.
 35. The method according to claim 33 further comprising after thedetecting of the at least one indication, verifying whether the noise ispresent based on at least one symbol of an unfiltered version of theuseful signal.
 36. The method according to claim 35 wherein theverifying comprises a direct Fourier transform processing on the atleast one symbol, and an analyzing of a power of each carrier.
 37. Themethod according to claim 35 wherein the verifying is carried out on atleast one symbol of the preamble sequence.
 38. The method according toclaim 37 further comprising processing the remaining part of each frame;and wherein when the noise is not present, the processing of theremaining part of a respective frame is carried out on the unfilteredversion of the useful signal.
 39. The method according to claim 38wherein each symbol comprises modulation coefficients respectivelyassociated with the sub-set of carriers; and wherein when the noise ispresent, the processing of the remaining part comprising direct Fouriertransform processing, de-mapping processing supplying a value of arespective modulation coefficient for each carrier, and determining aconfidence indicator of a value for each modulation coefficient and aforcing to zero of the confidence indicators of the modulationcoefficients associated with carriers whose frequencies correspond tothose of the noise.
 40. The method according to claim 39 wherein thevalues of the modulation coefficients are frozen during the processingof the remaining part of the frame.
 41. The method according to claim 29wherein the useful signal is a signal modulated according to anOrthogonal frequency-division multiplexing (OFDM) modulation.
 42. Themethod according to claim 29 wherein the transmission channel is anelectrical power line; and wherein the useful signal is a signal carriedby power line communications.
 43. A receiver comprising: an input stageto be coupled to a transmission channel and configured to deliver ananalog signal from the transmission channel, the analog signalcomprising a useful signal modulated on a sub-set of carriers from aplurality of available carriers, the useful signal carrying at least oneframe of symbols according to a frame structure, the useful signalsusceptible to noise; an analog-to-digital conversion stage configuredto perform an analog/digital conversion of the analog signal into adigital signal; and a processing stage comprising a filter configured todetermine, in a time domain, a number of coefficients of a predictivefilter from an autoregressive model of the digital signal, and apply adigital finite impulse response filter with coefficients based upon thenumber of coefficients to provide a filtered digital signal, and adetector configured to detect at least one indication for at least onelocation in the frame structure to be identified and based upon thefiltered digital signal and a reference signal.
 44. The receiveraccording to claim 43 wherein each symbol comprises a cyclical prefix;and wherein a number of the coefficients of the filtering of the digitalsignal is less than or equal to half of a sample length of the cyclicalprefix.
 45. The receiver according to claim 43 wherein said filter isconfigured to calculate the coefficients of the predictive filter at aprocessing frequency in a range between 2 and 5 times of a thresholdfrequency of the digital signal.
 46. The receiver according to claim 43wherein said filter is configured to: organize of samples of the digitalsignal into successive groups of samples; determine a current block ofcoefficients using a current group of samples; and apply to the currentgroup of the finite impulse response filter having the current block ofcoefficients so as to obtain a filtered current group of samples. 47.The receiver according to claim 43 wherein each frame comprises apreamble sequence comprising known symbols, and a remaining partsubsequent to the preamble sequence; and wherein said processing stageis configured to deliver the digital signal to said filter so that saiddetector detects the at least one indication within the preamblesequence of the at least one frame.
 48. The receiver according to claim47 wherein said processing stage is configured to deliver the digitalsignal to said filter so that said detector detects the at least oneindication within the preamble sequence of each and every frame.
 49. Thereceiver according to claim 47 further comprising a verification circuitconfigured to, after the detecting of the at least one indication,verify whether the noise is present based on at least one symbol of anunfiltered version of the useful signal.
 50. The receiver according toclaim 49 wherein said verification circuit comprises: a direct Fouriertransform stage configured to perform a direct Fourier transformprocessing on the at least one symbol; and an analyzer configured tocarry out an analysis of a power of each carrier.
 51. The receiveraccording to claim 49 wherein said verification circuit is configured tocarry out the verification on at least one symbol of the preamblesequence.
 52. The receiver according to claim 47 wherein said processingstage is configured to process the remaining part of each frame; andwherein the processing of the remaining part of a respective frame iscarried out on the unfiltered version of the useful signal
 53. Thereceiver according to claim 51 wherein each symbol comprises modulationcoefficients respectively associated with the sub-set of carriers; andwherein said processing stage is configured to, when the noise ispresent, process the remaining part of each frame, the processing of theremaining part comprising direct Fourier transform processing,de-mapping processing supplying a value of a respective modulationcoefficient for each carrier, and determining a confidence indicator ofa value for each modulation coefficient and a forcing to zero of theconfidence indicators of the modulation coefficients associated withcarriers whose frequencies correspond to those of the noise.
 54. Thereceiver according to claim 53 wherein said processing stage isconfigured to disable calculating of the coefficients of said filterduring the processing of the remaining part of the frame.
 55. Thereceiver according to claim 43 wherein the useful signal is a signalmodulated according to an Orthogonal frequency-division multiplexing(OFDM) modulation.
 56. The receiver according to claim 43 wherein thetransmission channel is an electrical power line; and wherein the usefulsignal is a signal carried by power line communications.
 57. A receivercomprising: an input stage configured to receive an electrical powerline analog signal comprising a useful signal modulated on a sub-set ofcarriers from a plurality of available carriers, the useful signalcarrying at least one frame of symbols according to a frame structure,the useful signal susceptible to noise; an analog-to-digital conversionstage configured to perform an analog/digital conversion of theelectrical power line analog signal into a digital signal; and aprocessing stage comprising a filter configured to determine a number ofcoefficients of a predictive filter from the digital signal, and apply adigital finite impulse response filter with coefficients based upon thenumber of coefficients to provide a filtered digital signal, and adetector configured to detect at least one indication for at least onelocation in the frame structure to be identified and based upon thefiltered digital signal and a reference signal.
 58. The receiveraccording to claim 57 wherein each symbol comprises a cyclical prefix;and wherein a number of the coefficients of the filtering of the digitalsignal is less than or equal to half of a sample length of the cyclicalprefix.
 59. The receiver according to claim 57 wherein said filter isconfigured to calculate the coefficients of the predictive filter at aprocessing frequency in a range between 2 and 5 times of a thresholdfrequency of the digital signal.
 60. The receiver according to claim 57wherein said filter is configured to: organize of samples of the digitalsignal into successive groups of samples; determine a current block ofcoefficients using a current group of samples; and apply to the currentgroup of the finite impulse response filter having the current block ofcoefficients so as to obtain a filtered current group of samples. 61.The receiver according to claim 57 wherein each frame comprises apreamble sequence comprising known symbols, and a remaining partsubsequent to the preamble sequence; and wherein said processing stageis configured to deliver the digital signal to said filter so that saiddetector detects the at least one indication within the preamblesequence of the at least one frame.